Design and Simulation of Microstrip Patch Antenna Using Ads

RF Propagation, Antennas, and Regulatory Requirements

Shahin Farahani , in ZigBee Wireless Networks and Transceivers, 2008

5.9.8.6 Patch Antennas

Patch antennas come in various shapes and sizes and consist of a patch of metal directly above a ground plane. Figure 5.27 shows an example of a patch antenna. The main disadvantage of these antennas is their relatively large size compared to other types of antennas. For example, some patch antennas are approximately half a wavelength on each side. The polarization can be either circular or linear depending on the design of the patch. In a patch antenna, most of the propagation is above the ground plane and can have high directional gain.

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Antennas and transmission lines

Alan Bensky , in Short-range Wireless Communication(Third Edition), 2019

3.2.7.2 Metamaterial dielectric substrates

Patch and loop antennas were designed with metamaterial dielectric substrates [10]. The aim here was to significantly decrease the physical size at the operating frequency. A loop antenna with approximately one wavelength perimeter that resonated in air at 2.58 GHz was mounted over a stack of pc boards on which were etched groups of split ring resonators as shown in Fig. 3.12. The resonant frequency with the metamaterial substrate was reduced by 23 percent, which indicates the size reduction. A greater decrease of resonant frequency was obtained when the split ring resonators were included in the same plane as the loop, both within the loop and outside it, resulting in a size reduction of 38 percent. A dielectric substrate similar to that shown in Fig. 3.12 was used for a patch antenna, where a ground plane larger than the size of the patch was placed at the bottom side of the substrate. The resonant frequency of the antenna with the metamaterial substrate was reduced somewhat more than what would be expected from the natural pc board substrate alone. Reduction of size of the patch and the loop antennas was accompanied by decreased bandwidth. In both cases, bandwidth was improved by adding nondriven antenna elements with slightly higher resonant frequencies on the opposite side of the dielectric from the driven element.

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Microwave Active Circuits and Integrated Antennas

William R. Deal , ... Tatsuo Itoh , in The Electrical Engineering Handbook, 2005

11.4.1 Microstrip Patch Antenna

The patch antenna has several desirable qualities, including a broadside radiation pattern that allows it to be integrated into two-dimensional arrays. The antenna is also low profile and low cost, has good conformability, and has ease of manufacturing. It is readily integrated with microstrip or coaxial probe feeding. Multilayer schemes have been used for other types of feeding, including CPW and strip line. With microstrip feeding, it is also a relatively simple task to implement either linear or circular polarization excitation of the antenna.

The patch antenna, shown in Figure 11.14 with microstrip feeding, is one of the most widely used planar antennas. Feeding is extremely important with the patch antenna, and it contributes to bandwidth, crosspolarization levels, and ripple. Microstrip-fed patches have very narrow bandwidths, almost invariably less than 5%. Other feed mechanisms have been used to increased bandwidth, including proximity coupling and aperture coupling, both of which require multilayer fabrication. A review of this technology is discussed in Pozar (1992). Alternatively, matched bandwidth of the antenna can be increased by making the antenna substrate electrically thicker, effectively lowering the Q-factor of the antenna cavity for increased bandwidth. High levels of TM₀ surface waves, however, can result and therefore reduce the radiation efficiency as well as degrade the radiation pattern if the surface wave generates radiation (which can occur at the edge of finite ground antennas). The problem of electrically thick substrate is also a common one for high-frequency antennas on high-permittivity substrates, and high amounts of TM surface waves can result.

Returning to Figure 11.14, the microstrip feed is inset into the antenna a distance x to obtain an input match. The dimension b is chosen so that the cavity formed by the conductor on the top plane of the structure is resonant. This causes radiation at the two edges of the antenna, as shown by the fringing fields in the diagram. A simple and intuitive technique for modeling this antenna is the transmission line model. This model provides a reasonable estimate for the resonant frequency and a fairly accurate estimate of the input impedance close to resonance. Due to the narrow bandwidth of the patch antenna, it is typically not accurate enough to guarantee first-pass design success. The bandwidth, however, does provide a useful starting point as well as useful insight into the operation of the antenna.

A cross section of the patch antenna is shown in Figure 11.15. In this model, it is assumed that the patch antenna consists of a perfect magnetic conductor (PMC) walls on the sides of the patch antenna, giving rise to standing wave type modes inside the patch antenna cavity. The total length of the cavity is the length of the patch antenna (dimension b) and an effective length at each edge due to the microstrip open-end effect. The fundamental resonance of the cavity formed by the microstrip patch antenna will occur at the frequency where the total effective length of the patch antenna, b + 2Δl_oc , is equal to one-half a guided wavelength in the microstrip cavity. The equation representing this concept is as follows:

(11.34) $f_{r} = \frac{c}{2 {\sqrt{ɛ}}_{r e}} \frac{1}{b + 2 Δ l_{o c}} .$

Note that c is the speed of light in a vacuum (c = 3*10⁸ m/s), and ε_re is the effective permittivity of the microstrip. In terms of the features in Figure 11.14, an approximate expression for the effective permittivity is as written here:

(11.35) $ɛ_{r e} = \frac{ɛ_{r} + 1}{2} + \frac{ɛ_{r} - 1}{2} {(1 + \frac{10 t}{a})}^{- \frac{1}{2}} .$

It is also necessary to have an estimate of the effective length due to the fringing effects. The following is a commonly used formula for the effective length of the fringing field:

(11.36) $\frac{Δ l_{o c}}{t} = 0.412 \frac{(ɛ_{r e} + 0.3) (a / t + 0.264)}{(ɛ_{r e} - 0.258) (a / t + 0.813)} .$

By using these equations together, dimensions of the patch antenna can be chosen to achieve a particular resonant frequency. The accuracy of the model is typically a few percent. Because the bandwidth of a microstrip-fed patch is on the same order, this may not be accurate enough for first pass design.

It is also desirable to have an estimate of the input impedance of the antenna. The simplest way for estimating this is the transmission line model for the patch antenna. In this case, the antenna is modeled as two radiating slots of width Δl_oc and length a separated by a microstrip transmission line with dimensions corresponding to the dimensions of the patch antenna. Note that the feeding can be placed at one end of the antenna or at some point a distance x inside the patch, either by the use of an inset feed or a coaxial probe. The equivalent structure for interior feeding is shown in Figure 11.16. At resonance, the impedance of the radiating slot will be pure real. To first-order for a ≪ λ₀ (which will be true on high-permittivity substrates), the radiation resistance of each slot may be approximated as:

(11.37) $R_{a} = \frac{90 λ_{0}^{2}}{a^{2}} .$

Equation 11.37 provides a reasonable first-order estimate of the input impedance of the patch antenna near resonance. Note that there are many other simple models that also provide a first-order estimate of the input impedance. Often, the designer must either fabricate and perform measurements or obtain a full-wave solution to the structure using an electromagnetic (EM) simulator to achieve accurate design data for the antenna. This simple model, however, provides a useful starting place for the design.

Because the antenna's length determines the resonant frequency of the patch, higher antenna resonances will coincide with frequencies that are multiples of the fundamental resonance. If active circuitry, which may generate harmonic frequencies, is integrated with the patch antenna, harmonic radiation leading to co-site interference may occur. A circular geometry patch antenna may be used to reduce this problem. In this case, higher resonances of the antenna will be determined by circular harmonics (Bessel functions) and can be designed to occur away from circuit harmonics. A plot of the input impedance of one particular type of circular geometry patch, the circular segment patch antenna, is shown in Figure 11.17 along with its geometry. By looking at the frequency scale of the plot, it is apparent that higher resonances do not correspond to harmonic frequency of active devices that may be integrated with the antenna. Note that the input impedance was obtained by full-wave analysis. Measured radiation patterns of these types of antenna are comparable to that of a standard rectangular geometry patch antenna. In addition, circular geometry patch antennas are often more compact than rectangular geometry patch antennas. Note that the antenna is fabricated on a standard RT/Duroid of permittivity 2.33 and a thickness of 31 mils. A 120° sector of the antenna has been removed for optimal impedance. A microstrip feed is placed 30° from the edge of the voided sector. The radius of the antenna is 740 mils.

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Magnetoelectric composites for miniature antennas

G. Yang , N.X. Sun , in Composite Magnetoelectrics, 2015

10.6.7 Bulk composites

A patch antenna with the dimension of 10 × 8 cm ² is printed on the four-layer magneto-dielectric materials with thicknesses of 2 cm and illustrated in Figure 10.36. The size of the ground plane is 20 × 20 cm². The return loss and radiation patterns are shown in Figure 10.37(a) and (b), respectively. The resonant frequency of the fabricated antenna is about 277 MHz, and it provides a wide bandwidth of about BW = 3.2%. The size of the antenna is around with a miniaturization factor of 5.4. The directivity of the antenna is D ₀ = 2.9 dB, it has a front-to-back ratio of 1.3 dB with a ground plane size of 0.18λ ₀ × 0.18λ ₀, and the antenna efficiency is about e _r = 67%. If a magneto-dielectric material with lower magnetic loss tangent of about 0.01 is used, the efficiency is increased to 82% while the bandwidth is decreased to BW = 2.8% (Mosallaei & Sarabandi, 2004).

It should be noted that to achieve the same miniaturization factor utilizing only a dielectric material (μ _r = 1), the relative permittivity would be 23.7. This reduces the bandwidth to about BW = 0.5%, which is shown in Figure 10.37(a). The efficiency in this case for a dielectric loss tangent of 0.001 is about e _r = 64%. Therefore, utilizing the magneto-dielectric meta-substrate, one can offer a miniaturized wideband planar antenna with high efficiency. The antenna bandwidth for the proposed magneto-dielectric substrate is about six times higher than that of the dielectric substrate.

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Array Antenna Elements

NICHOLAS FOURIKIS , in Advanced Array Systems, Applications and RF Technologies, 2000

3.6.3.1 The Electromagnetically Coupled (EMC) Patch

The EMC patch antenna, shown in Figure 3.17a, consists of a radiating patch and a feeding patch separated by a layer of foam. As can be seen, the designer now has several degrees of freedom, e.g. the diameters of the radiating and feeding patches, D _r and D_f , the relative permittivity of the substrates on which the two patches are realized, and the relative permittivity and height of the foam. Additionally, there are no transmission lines feeding the EMC which is now fed at points A, B, C, or D. With this arrangement the radiating patch is protected from the weather and can be conformal to the skin of the platform. The feed has a wider bandwidth than its conventional counterpart [129] and this attractive feature is directly attributed to its dual-resonant structure. The EMC patch is easy to integrate with an MMIC-based T/R module that uses a high relative permittivity substrate such as GaAs. This important attribute of the EMC patch resolves the difficulty we have allured to, i. e. the integration of the antenna and to the MMIC-based T/R module.

An EMC patch having a 13% bandwidth and a maximum cross-polarization level of −30 dB at L-band is reported [130] and design guidelines for the EMC patch are given. Similarly, a broadband EMC patch operating at 1.67 GHz has been reported [131]. It has a 1.7 VSWR bandwidth of 14.4% and its cross-polarization level is better than −27 dB over the same frequency range, and comparable performance was obtained by another EMC patch operating at S-band [132]. These are impressive characteristics for low-cost antennas that have all the other desirable properties.

These experimental results are supported by recent theoretical work related to ECM patches [133,134]. Reference [135] is a recent, useful reference for antenna elements normally considered for phased array applications.

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Embroidered antennas for communication systems

Z. Wang , ... A. Kiourti , in Electronic Textiles, 2015

10.4.3.1 Embroidered textile patch antenna

First, an embroidered textile patch antenna was fabricated on a polymer substrate ( ɛ _r = 4.2 and tan δ = 0.01). Next, the antenna was measured on planar and cylindrical surfaces (see Figure 10.17). As shown in Figure 10.17a, the measured resonance frequency of the planar textile patch antenna was 2.2 GHz, agreeing with that of its simulated copper counterpart. The measured realized gain was 5.6 dBi, which is only 0.3 dB lower than that of the copper patch antenna. The measured patterns were also in agreement with simulations. We further note that this remarkable RF performance of the textile antenna does not degrade after repetitive flexing (more than 20 times).

To further evaluate the RF performance of the E-fiber antenna, the latter was mounted on a metallic cylinder (diameter = 80 mm). As shown in Figure 10.17b, the measured reflection coefficient and radiation patterns of the textile patch antenna were in good agreement with the simulations of the equivalent copper patch antenna. The realized gain was only 1 dB lower than simulation. Further, as compared to the flat configuration, the textile antenna had a lower resonance frequency of 2.06 GHz and a reduced gain of 3.0 dB. The frequency detuning is due to the 13% elongation of the patch dimension in the H-plane (Wang et al., 2012b). However, the gain reduction is primarily because of the curvature and higher resistance of the textile's surface. The latter was due to the stretching of the E-fiber threads. Nevertheless, these results clearly demonstrate the remarkable RF and mechanical performance of the E-fiber antenna.

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Metasurfaces

Tatjana Gric , Ortwin Hess , in Phenomena of Optical Metamaterials, 2019

Coupling Between Free Space and Surface Waves

The study [19] reports on the revolutionary work on using metasurfaces to control guided waves in the microwave spectral range. Thus, the concept of holography to design high-impedance surfaces converting a given surface wave into a freely propagating wave possessing desired far-field radiation pattern and polarization was introduced in Ref. [19]. The (high) impedance surface is basically a hologram, with former being the interference pattern between a reference beam and an object beam and carries information of the phase, amplitude, and polarization of the desired object beam. The object beam is reconstructed when the reference beam impinges on the hologram. The reference beam in the form of a surface wave, E _surf, is produced by a source antenna and the object beam is the desired wave, E _rad, propagating in the half space above the surface (Fig. 3A ) [19]; to create the microwave holograms consisting of a square lattice of dissimilar subwavelength conductive patches on a metal-grounded dielectric substrate, the interference pattern produced by the two waves is used. Both scalar and tensor impedance surfaces have been experimentally shown.

Surface impedance provides a fertile basis to characterize the properties of the metasurface. It is defined as the ratio between the electric and magnetic fields near the surface. The surface impedance is Z(x, y) = E _x(x, y)/H _y(x, y) for transverse magnetic (TM) waves (i.e., magnetic field transverse to the propagation direction) that propagate in the x-direction. The surface magnetic field is proportional to the surface current, which is provided by the electromagnetic source. For example, a cylindrical distribution of surface current is produced by a monopole antenna. The impedance surface stands for the translation of this surface current to a distribution of electromagnetic waves on the surface, which matches the desired radiative wave.

The goal of usage of the square patch antennas ( Fig. 3B) was to create scalar impedance surfaces and the square patches with an additional slice were used for tensor impedance surfaces [19]. There are three degrees of freedom to obtain the required performance of the antenna design: the slice width, its orientation angle, and the gap between neighboring square patches in order to control three independent terms in the impedance tensor, Z _xx, Z _xy = Z _yx, and Z _yy. In the case of scalar impedance surfaces, the following procedures are needed in order to determine the value of surface impedance of patch antennas:

(i): Calculating dispersion relation of surface waves propagating on a 2D periodic array of patch antennas. Explicitly, Bloch boundary conditions are applied to a unit cell of the impedance surface, and eigensurface wave modes and their eigenwave vectors are determined for a range of frequencies.
(ii): Calculating the surface impedance for a given operation frequency ω ₀, Z(ω ₀) = ∫_{unit cell}(E _x/H _y)dxdy.

It is possible to create a library relating the surface impedances and patch antenna geometries by repeating the process explained above for patch antennas having different sizes.

One might define the distribution of surface impedance Z(x, y) by means of the following holographic technique. It is worthwhile mentioning that the required surface impedance is $Z (x, y) = j \{X + M R_{e} [(E_{rad, x}, E_{rad, y}) (\begin{array}{c} J_{surf, x} \\ J_{surf, y} \end{array})]\}$ in the case of a scalar impedance surface, with a surface current J _surf (x, y) produced by the electromagnetic source and the object far-field radiation E _rad(x, y, z). In the case of a tensor impedance surface, we have

$Z (x, y) = j (\begin{array}{c} X & 0 \\ 0 & X \end{array}) + j \frac{M}{2} Im [(\begin{array}{c} E_{rad, x} \\ E_{rad, y} \end{array}) (J_{surf, x}, J_{surf, y}) - (\begin{array}{c} J_{surf, x} \\ J_{surf, y} \end{array}) (E_{rad, x}, E_{rad, y})] .$

In the above two equations, X represents the average impedance value, and M spans the entire available impedance range. By means of the holographic technique and the library of patch antennas, a scalar impedance surface scattering the current generated by a monopolar antenna into a linearly polarized plane-wave propagating along 60° from the surface normal (Fig. 3C and E) was presented in Ref. [19]. The surface current has a cylindrical distribution and can be described by $J_{surf} = \frac{1}{r^{2}} exp (- {jk}_{0} n_{s} r) (x, y, 0)$ , where r = (x ² + y ²)^1/2, k ₀ is the free-space wave vector, and n _s is the effective index of the surface current. It is worthwhile mentioning that the former parameter is assumed to be a constant and is a function of the thickness and materials of the dielectric spacing layer between the metal patches and the metallic ground. A tensor impedance surface converting the current generated by a monopolar antenna to a circularly polarized far-field radiation propagating along 45° direction (Fig. 3D and F) was also experimentally demonstrated in Ref. [19].

The same holographic principle to demonstrate metasurfaces with modulated surface impedance was proposed in Refs. [20, 21]. The square patch antennas of different sizes to create a spiral distribution of surface impedance converting a surface current produced by a monopolar antenna to a collimated right-handed circularly polarized far-field radiation (Fig. 4) were used. It was demonstrated experimentally [22] that width-modulated microstrip lines patterned on a grounded dielectric slab present a sinusoidally modulated surface impedance and generate leaky waves (Fig. 5). A holographic metasurface that detects optical vortex beams with specific orbital angular momentum (OAM) [23] is shown in Fig. 6. The nanostructured binary holograms shown in the left panel of Fig. 6A were produced by interfering a converging surface plasmon wave with an incident optical vortex beam. The simulated results in Fig. 6A show that a converging surface plasmon wave is generated merely when an optical vortex beam with the correct OAM is scattered by the hologram. Experimental results in Fig. 6B demonstrate that a hologram can distinguish an optical vortex beam with OAM of − 1 from optical vortex beams with other values of OAM.

The major goals of coupling an incident wave from free space into a surface wave with high efficiency are to suppress the reflection of the incident wave on the device surface and to escape from decoupling of the surface wave back into free space. A few strategies were devised to address these challenges in Refs. [24–26]: (1) the entire surface of the metasurface coupler is considered to be impedance matched with free space in order to reduce direct reflection; (2) the lateral effective wave vector provided by the metasurface is designed to be sufficiently large with the intention of exciting a surface wave with a wave vector larger than the free-space wave vector; it is possible to extend the degree of the surface wave evanescence by its interaction with the gradient metasurface, preventing decoupling of the waveback to the free space; (3) the impedance mismatch between the super cells of the metasurface coupler is reduced in order to prevent scattering of the surface wave. The former approaches have provided a fertile ground to demonstrate coupling of an incident wave from free space into a surface wave with efficiencies of ~ 94% in simulations and ~ 73% in experiments using microwaves [26].

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Latest developments in the field of textile antennas

L. Vallozzi , ... H. Rogier , in Smart Textiles and their Applications, 2016

26.4.2.1 2.45 GHz

Hertleer et al. [24] introduced a wearable patch antenna for operation in the [2.4, 2.4835] GHz band, made of a shock-absorbing, fire-retardant foam substrate and suitable for rescue worker applications. The antenna employs a simple topology, consisting of a truncated-corner rectangular patch (see Fig. 26.6), with coaxial feeding on the patch diagonal, allowing the excitation of two orthogonal modes that yield nearly circular polarization. The conductive parts of the antenna are composed of commercially available electrotextile materials, with very high conductivity, in particular "ShieldIt™" for the radiating patch and "Flectron^®" for the ground plane. The proposed antenna was first designed by means of an optimization performed by a full-wave EM solver, then prototyped and tested in different operating conditions. In particular, the authors showed, through both measurements and simulations, how the proximity of a human arm over which the antenna was bent affects its performance by producing a resonance frequency shift. However, thanks to the design's relatively large −10 dB impedance bandwidth, the antenna still meets the design requirements when bent around a human arm.

In 2008, using the same protective foam as a substrate, Vallozzi et al. [11] proposed a 2.45 GHz patch antenna with dual polarization, allowing to implement polarization diversity using a single, compact, wearable antenna. The antenna, shown in Fig. 26.7, employs a simple nearly square topology with a small slot in the center and with two coaxial feeds positioned symmetrically on the two patch diagonals. This allows the excitation of two orthogonal, linearly polarized waves that can simultaneously transmit/receive two independent radio waves for implementation of polarization diversity. Antenna design and optimization were first performed with the aid of a full-wave EM solver, and then a prototype was constructed and its performance tested by measurements. In addition to a free-space situation, measurements were also performed on a human body in order to verify the antenna's resilience to the presence of a human body. In both cases, the measured reflection coefficients meet the design requirements (|S ₁₁| < –10 dB; |S ₂₂| < –10 dB), with the resonance frequency undergoing only a very small shift in the on-body situation. Measured gain patterns also remain almost unchanged in the on-body case, with respect to the free-space situation, with the maximum value being about 6 dBi in the broadside direction, which is more than sufficient to establish a reliable off-body communication link. Elliptical polarization was obtained in an on-body situation, versus nearly linear polarization in a free-space scenario. In the on-body case, however, the two polarization ellipses remained quasi-orthogonal to each other, ensuring the independence of the two signals needed to obtain diversity gain.

To apply diversity in off-body communication, two such antennas can be integrated into a wearable textile system, worn on the front and back side of a human subject, realizing a fourth-order receive diversity communication link by combining both pattern and polarization diversity [25]. By means of a realistic measurement campaign, it was shown that the proposed diversity system achieves a dramatic improvement in terms of received BERs in an off-body wireless communication link between a fixed transmitting base station and a receiving subject equipped with such an antenna system, moving into a typical indoor multipath environment.

More recently, a novel and promising technology called substrate integrated waveguide (SIW), an already well-known fabrication technology for rigid printed circuit boards, was for the first time applied by Moro et al. to develop a wearable textile antenna for operation in the 2.45 GHz ISM band [26]. Such a kind of patch antenna answers to important requirements in off-body communications, such as suppression of undesired surface waves and a high level of shielding from the human body, even with a very small ground plane, high directivity, and front-to-back ratio, as well as performance stability. The structure consists of a cavity-backed slot antenna on a flexible protective foam substrate, with electrotextiles (Flectron) metallization on both sides. The top layer consists of a rectangular conductive layer with a dog-bone-shaped slot, representing the radiating element, while on the bottom layer a 50 Ω grounded coplanar waveguide represents the feeding line. A rectangular cavity is formed inside the antenna substrate, by using eyelets as metallized holes, with an appropriate spacing distance between each other. All geometrical parameters were optimized by means of a commercial full-wave EM solver, and after that the antenna was prototyped by a low-cost production technique. An experimental verification of the antenna performance in free space showed that the antenna exhibits a reflection coefficient lower than −10 dB in a bandwidth of 165 MHz, including the complete 2.45 GHz ISM band, a maximum gain of 3.21 dBi at 2.45 GHz, and a radiation efficiency of 68%. Experiments were repeated with the antenna integrated on the back side of a body-worn firefighter jacket, showing that the performance deviates only slightly from the free-space state, with the maximum gain increasing to 4.9 dBi, owing to reflections by the human body. Effects of bending were also verified by simulations, resulting in a very small increase of resonance frequency for a bending radius of 10 cm, which does not compromise the overall performance in the operation band.

Later, the SIW cavity-backed slot antenna was used by Lemey et al. [27] as a starting point for the development of a novel energy-harvesting platform, obtained by solar cells and dedicated flexible circuitry, compactly integrated on the top and back surface of the SIW antenna structure, as depicted in Fig. 26.8. More specifically, two a-Si:H-solar cells were applied on the top side of the SIW antenna, while the necessary circuitry for the management of the output DC power from the solar harvesters was integrated on the back side and connected to the solar cells by means of wires routed through the eyelet holes.

The integrated circuits were composed of a central power management system (CPMS) and a low power system (LPS), deployed on a flexible polyimide substrate on a ground plane, glued on the back side of the SIW. More details of the implementation are described in Ref.[27].

Two other interesting compact structures for the implementation of wearable textile patch antennas operating at 2.45 GHz were proposed and studied by Liu et al. [28]. These antennas both reduced the size of a conventional structure by half, using electrical or magnetic symmetries. In particular, the first antenna, called the quarter-wave patch, is built up by using half of a rectangular patch, by placing a shorting wall providing electrical symmetry. The second antenna, the half-mode cavity, originates from a half-mode substrate-integrated cavity, where an open aperture placed on the substrate's symmetry plane enables to only use one-half of the entire cavity. The authors first developed an analytical analysis of both structures. Next, they built accurate EM simulation models, reproducing the features of their real textile implementation. Both antennas are made out of a low-loss nonabsorbent microwave radome foam PF-4 (thickness h = 3.2 mm and Ɛ_r = 1.06), with a ground plane with size 10 × 10 cm, and a half-square top conducting layer consisting of silver-coated fabric NCS95R-CR. The shorting sides (one on the symmetry plane for the quarter-wave antenna, and three on the peripheral sides for the half-mode) are realized by linear embroidery with a conductive thread, consisting of stitches with 1-mm spacing and a total of 5 passes. Through simulations, it was found that seam compression caused by stitching produces a resonance frequency shift with respect to the ideal planar case. Hence, such an effect needs to be included in the model to correctly predict resonance frequency and performance. Real prototypes of the two antennas were built and their performance was experimentally assessed obtaining operational bandwidths of 300 and 130 MHz, and maximum gains of 5.3 and 5.1 dBi, for the quarter-wave patch and half-mode cavity, respectively. Moreover, the effects of human body proximity on the performance parameters appeared to be indistinguishable, proving the good isolation properties of the ground plane.

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Terahertz integrated devices and systems

T. Ouchi , in Handbook of Terahertz Technology for Imaging, Sensing and Communications, 2013

14.2.2 Design of RTD oscillators

A novel type of resonant structure for RTD oscillators with a patch antenna can realize higher efficiencies ( Sekiguchi et al., 2010). The device structures of InGaAs/InAlAs RTD oscillators with a patch antenna are shown in Fig. 14.5. An RTD post, buried within a dielectric, is sandwiched between the patch and the ground plane on InP substrates. The oscillation frequency is decided by the resonant length L of the patch. The resonant length L is determined by a value of λ/2√ε _r (Kraus and Marhefka, 2001), where λ is the oscillation wavelength in a vacuum and ε, is the relative permittivity of the dielectric. The input resistance of the patch can be tuned, by shifting the RTD post over a distance x away from the center of the patch, proportionally to sin2(πx/L) (Kumar and Ray, 2003). Bias voltage on the RTD post is supplied at the center null point of the patch. A parallel resistance is put between the bias supply lines in order to prevent parasitic oscillation due to return pass of the bias supply lines.

The oscillation output of the RTD oscillators is maximized when the impedance of the antenna is matched to the RTDs. Although the patch antenna needs high gain RTDs due to its large load, the antenna that has several tens of ohms can satisfy the matching condition. The patch antenna of planar structures on substrates is easy to fabricate even for the array scheme. Since almost all fields emit toward the upper side of the substrate, dielectric lenses are not essential. As the oscillation output of the slot antenna fabricated by Suzuki et al. (2010) emits mainly into the substrate, the lens on the opposite surface is required to avoid substrate modes.

In the actual device, the RTD post of 2 to 4 μm-diameter on the InP substrate is formed by electron beam lithography and dry-etching techniques. An approximately 3 μm-thick layer of BCB is used to provide the magnitude of impedance of the patch antenna at about 50 Ω. Ti/Pd/Au metal in contact with the RTD post and Ti/Au metal in contact with the n-type InP substrate are adopted as the patch and the ground plane, respectively.

InGaAs/InAlAs triple-barrier RTDs were used as a gain structure for THz frequency range (Asada, 2001), which suppresses the broadening of sub-bands at quantum wells. Therefore, these types of RTDs enable resonant sharpness to be narrower compared with double-barrier RTDs (Nakagawa et al, 1986). As a result, large negative differential conductance is expected against small peak current density. The triple-barrier RTD is designed to get negative differential conductance similar to double-barrier RTDs. The epitaxial layers consist of (from the top down), n⁺InGaAs (Si = 1 × 10¹⁹ cm³, 100 nm), n-InGaAs (Si = 2 × 10¹⁸ cm³, 50 nm), InGaAs spacer (undoped, 5 nm), AlAs barrier (undoped, 1.3 nm), InGaAs quantum well (undoped, 7.6 nm), InAlAs barrier (undoped, 2.6 nm), InGaAs quantum well (undoped, 5.6 nm), AlAs barrier (undoped, 1.3 nm), InGaAs spacer (undoped, 5 nm), n-InGaAs (Si = 2 × 10¹⁸ cm³, 50 nm), and n + InGaAs (Si = 1 × 10¹⁹ cm³, 3400 nm) on the n-type InP substrate. The InGaAs and InAlAs layers are lattice-matched to InP substrates.

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POLYMERIC STRUCTURES OPTIMIZED FOR ORGANIC PASSIVE ELECTRONIC COMPONENTS

Sulaiman Khalifeh , in Polymers in Organic Electronics, 2020

7.8 ORGANIC ANTENNAS

The organic/polymeric antennas represent the seventh type of organic passive electronic components, which are polymer-based. The electric power is converted to radio waves with a part having high flexibility, printability, operating with high frequencies, and light weight. These antennas are applied in communication, human skin contact, and radio frequency identification. The optimized polymers/materials include polyurethane, poly(ethylene terephthalate), silicones, liquid crystal polymers, polyimides film (including commercial-grade Kapton® ^194, ¹⁹⁵ ), acrylonitrile-butadiene-styrene, poly(vinyl chloride), epoxy-glass fire retardant grade-4, poly(ethylene dioxythiophene), polydimethylsiloxane, polypyrrole, carbon nanotubes, polyamides, polytetrafluoroethylene, etc. According to their application and polymeric structure, the polymeric antennas can be classified as

1.: flexible polymer antenna on textiles
2.: polymer flexible-patch antenna
3.: polymer flexible-skin contact antenna
4.: polymer flexible-radio-frequency identification antenna. ^56, ⁹²

The flexible polymer antenna on textile is thin, flexible, lightweight, portable, mobile, and wearable antenna integrated with textiles to satisfy the following functions, such as monitoring, communication, energy harvesting, and storage. Here, textile dielectric or conductive nature is utilized to act as a flexible substrate. The conductive materials include copper, nickel, silver, etc. The optimized dielectric materials (polymers) include polyurethanes, polydimethylsiloxane, polyesters, polyimides, polyamides, etc. Several methods can be used for manufacturing polymer flexible antenna on textile, such as

1.: inkjet printing technique by which antenna is directly inkjet printed on one of three different substrates (polyimide PI (Kapton® ^194, ¹⁹⁵ ), polyurethane-coated stretchable textile, and pretreated polyesters/cotton textile). An example of the inkjet printable conductive inks is a silver nanoparticle dispersion (including the commercial-grade U5714® ¹⁸⁸ )
2.: carbon nanotubes and conductive polymer electronic ink for the textile technique represent process in which the electronic ink printed substrates act as flexible antenna. With this technique, a mixture of poly(3,4-ethylene dioxythiophene) doped with polystyrene sulfonate PEDOT:PSS with ethylene glycol can be used as electronic ink (known as "conductive polymer-based PEDOT-PSS ink"). Sodium dodecylbenzenesulfonate is used as a surfactant. Polyvinylphenol is used as a hydrophilic polymer. Single-walled carbon nanotube, multi-walled carbon nanotubes, and nitrogen-containing carbon nanotubes are also used
3.: deposition technique represents the process in which conductive material/polymer is deposited on textile or polymeric flexible substrate. Textiles coated on both sides with poly(vinyl chloride) are used as a flexible substrate for thin conductive film deposition. Additionally, polyamide polymers such as Kapton® ^194, ¹⁹⁵ can be chosen as reference substrate material. ^{61,93,95–96,135–136}

The second type of organic antennas, the polymer flexible-patch antenna (also called "polymer rectangular microstrip antenna)" represents a generation of polymeric radio antenna with a low profile, which can be mounted on a flexible flat surface. Optimizing the polymeric structure of this type depends mainly on the right selection of both conductive materials (metals) and textiles made from conductive coated materials such as conductive yarns of polyamide, thermoplastic polyurethane, polyurethane foam, aramid fabric, or cotton/polyester mixture fabric. ^97, ¹³¹

As the third type of organic antennas, the polymer flexible-skin contact antenna is selected as an example of an organic antenna based on thin and flexible film of conducting polymers. It is used for medical imaging, diagnosis, and treatment. As illustrated in Figure 7.12, ⁹⁸ this organic antenna can be placed in contact with human skin. Optimization of its polymeric structure depends on the use of polymeric layers, such as polyimide (e.g., Kapton® ^194, ¹⁹⁵ having a dielectric constant of 3.5) and other types of thermosetting layers acting as adhesives (including polyamide film having a dielectric constant of 3.1). In this case, the human skin acts as a substrate layer of 1.5 mm thickness, dielectric coefficient of 39, and a conductivity of 1.1 S/m. The antenna is fabricated from flexible copper, having a thickness of 250 µm. The polyamide layer can be used in the form of a laminated sheet, thin film, thick film, etc. All these types are flexible and can be constructed using adhesives. ^56, ^65, ⁹⁸

As the fourth type of organic antennas, the polymer flexible-radio-frequency identification antenna, which uses wireless technique to identify the predetermined objects, can be structured using optimized polymers, especially those having considerable transparency, flexibility, and ability to be applied by screen printing process such as poly(ethylene terephthalate), polyimide (e.g., Pyralux® ^78, ¹⁹⁶ or Kapton® ^194–195 ), and polyesters (e.g., Rogers RO4003® ²¹⁶ ). Such polymeric materials should be built on flexible polymer-based substrates. Note: poly(ethylene terephthalate) and poly(ethylene naphthalate) films are used for structuring these antennas by screen printing using copper or printed polymer ink as a conductive layer. It is called a clear near field communication tag. Poly(ethylene terephthalate) is desirable for such a structure because it is a transparent film with high flexibility. Poly(ethylene terephthalate) transparent film is used widely as a clear or transparent film in structuring near field communication tags as a flexible substrate (Figure 7.13 ¹⁹⁷ ). Both antenna and poly(ethylene terephthalate) film are protected on both sides with adhesive of thermosetting type. ^100, ¹⁰²

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Design and Simulation of Microstrip Patch Antenna Using Ads

Source: https://www.sciencedirect.com/topics/engineering/patch-antenna

Design and Simulation of Microstrip Patch Antenna Using Ads

RF Propagation, Antennas, and Regulatory Requirements

5.9.8.6 Patch Antennas

Antennas and transmission lines

3.2.7.2 Metamaterial dielectric substrates

Microwave Active Circuits and Integrated Antennas

11.4.1 Microstrip Patch Antenna

Magnetoelectric composites for miniature antennas

10.6.7 Bulk composites

Array Antenna Elements

3.6.3.1 The Electromagnetically Coupled (EMC) Patch

Embroidered antennas for communication systems

10.4.3.1 Embroidered textile patch antenna

Metasurfaces

Coupling Between Free Space and Surface Waves

Latest developments in the field of textile antennas

26.4.2.1 2.45 GHz

Terahertz integrated devices and systems

14.2.2 Design of RTD oscillators

POLYMERIC STRUCTURES OPTIMIZED FOR ORGANIC PASSIVE ELECTRONIC COMPONENTS

7.8 ORGANIC ANTENNAS

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